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2101d?bdc?06/04 features up to 2 gsps sampling rate power consumption: 4.6 w 500 mvpp differential 100 ? or single-ended 50 ? ( 2 %) analog inputs differential 100 ? or single-ended 50 ? clock inputs ecl or lvds output compatibility 50 ? differential outputs with common mode not dependent on temperature adc gain adjust sampling delay adjust offset control capability data ready output with asynchronous reset out-of-range output bit selectable decimation by 32 functions gray or binary selectable output data; nrz output mode pattern generator outp ut (for acquisition system monitoring) radiation tolerance oriented design (more than 100 krad (si) expected) cbga 152 cavity down hermetic package cbga package evaluation board tsev83102g0bgl companion device: dmux 8-/10- bit 1:4/1:8 2 gsps ts81102g0 performance 3.3 ghz full power input bandwidth (-3 db) gain flatness: 0.2 db (from dc up to 1.5 ghz) low input vswr: 1.2 max from dc to 2.5 ghz sfdr = -59 dbc; 7.6 effective bits at f s = 1.4 gsps, f in = 700 mhz [-1 dbfs] sfdr = -53 dbc; 7.1 effective bits at fs = 1.4 gsps, f in = 1950 mhz [-1 dbfs] sfdr = -54 dbc; 6.5 effective bits at f s = 2 gsps, f in = 2 ghz [-1 dbfs] low bit error rate (10 -12 ) at 2 gsps application direct rf down conversion wide band satellite receiver high-speed instrumentation high-speed acquisition systems high-energy physics automatic test equipment radar screening temperature range for packaged device: ? ?c? grade: 0 c < tc; tj < 90 c ? ?v? grade: -20 c < tc; tj < 110 c standard die flow (upon request) description the ts83102g0b is a monolithic 10-bit analog- to-digital converter, designed for digi- tizing wide bandwidth analog signals at very high sampling rates of up to 2 gsps. it uses an innovative architecture, including an on-chip sample and hold (s/h). the 3.3 ghz full power input bandwidth and band flatness performances enable the digitiz- ing of high if and large bandwidth signals. 10-bit 2 gsps adc ts83102g0b
2 ts83102g0b 2101d?bdc?06/04 figure 1. simplified block diagram functional description the ts83102g0b is a 10-bit 2 gsps adc. the device includes a front-end master/slave track and hold stage (sample and hold), followed by an analog encoding stage (analog quantizer), which outputs analog residues resulting from analog quantization. successive banks of latches regenerate the analog residues into logical levels before entering an error correction circuit and resynchronization stage, followed by 50 ? differential output buffers. the ts83102g0b works in a fully differential mode from analog inputs to digital outputs. a dif- ferential data ready output (dr/drb) is avail able to indicate when the outputs are valid and an asynchronous data ready reset ensures that the first digitized data corresponds to the first acquisition. the control pin b/gb (a11 of the cbga package) is provided to select either a binary or gray data output format. the gain control pin ga (r9 of the cbga package) is provided to adjust the adc gain transfer function. a sampling delay adjust function (sda) may be used to ease the interleaving of adcs. a pattern generator is integrated on the chip for debug or acquisition setup. this function is activated through the pgeb pin (a9 of the cbga package). an out-of-range bit (or/orb) indicates when the input overrides 0.5 vpp. a selectable decimation by 32 functions is al so available for enhanced testability coverage (a10 of the cbga package), along with the die junction temperature monitoring function. the ts83102g0b uses only vertical isolated npn transistors together with oxide isolated pol- ysilicon resistors, which allows enhanced radiation tolerance (over 100 krad (si) total dose expected tolerance). sample &hold clock generation logic block analog quantizer vin vinb cl k clk b pgeb b/gb drrb or orb d9 d9b d0 d0b ga dr drb 50 50 50 50 sda sda decb/ diode
3 ts83102g0b 2101d?bdc?06/04 specification note: absolute maximum ratings are short term limiting values (re ferenced to gnd = 0v), to be applied individually, while other parameters are within specified operating conditions. long exposu re to maximum ratings may affect device reliability. all inte- grated circuits have to be handled with appropriate care to avoid damage due to esd. damage caused by inappropriate handling or storage could range from performance degradation to complete failure. absolute maximum ratings parameter symbol comments value unit positive supply voltage v cc gnd to 6.0 v digital negative supply voltage d vee gnd to -5.7 v digital positive supply voltage v plusd gnd - 1.1 to 2.5 v negative supply voltage v ee gnd to -5.5 v maximum difference between negative supply voltages d vee to v ee 0.3 v analog input voltages v in or v inb -1.5 to 1.5 v maximum difference between vin and vinb v in - v inb -1.5 to 1.5 v clock input voltage v clk or v clkb -1 to 1 v maximum difference between vclk and vclkb v clk - v clkb -1 to 1 vpp static input voltage v d ga, sda -5 to 0.8 v digital input voltage v d sdaen, drrb, b/gb, pgeb, decb -5 to 0.8 v digital output voltage v o v plusd min operating -2.2 to v plusd max operating + 0.8 v junction temperature t j 130 c recommended condi tions of use parameter symbol comments min typ max unit positive supply voltage v cc 4.75 5 5.25 v positive digital supply voltage v plusd differential ecl output compatibility - 0.9 - 0.8 - 0.7 v lvds output compatib ility 1.375 1.45 1.525 v grounded (1) maximum operating vplusd 1.7 v negative supply voltages v ee , d vee - 5.25 - 5.0 - 4.75 v differential analog input voltage (full-scale) v in , v inb v in - v inb 50 ? differential or single-ended 113 450 125 500 137 550 mv mvpp clock input power level (ground common mode) p clk , p clkb 50 ? single-ended clock input or 100 ? differential clock (recommended) - 4 0 4 dbm
4 ts83102g0b 2101d?bdc?06/04 note: 1. adc performances are independent on v plusd common mode voltage and performances are guaranteed in the limits of the specified v plusd range (from -0.9v to 1.7v). operating temperature range commercial "c" grade industrial "v" grade 0c < t c ; t j < 90c -20c < t c ; t j < 110c c storage temperature tstg -65 to 150 c lead temperature tlead 300 c electrical operatin g characteristics v cc = 5v ; v plusd = 0v (unless otherwise specified). adc performances are independent of v plusd common mode voltage and performances are guaranteed within the limits of the specified v plusd range (from -0.9v to 1.7v); v ee = d vee = -5v; v in - v inb = 500 mvpp (full-scale single-ended or differential input); clock inputs differential driven; analog-input single-ended driven. parameter test level symbol min typ max unit resolution 10 bits power requirements positive supply voltage - analog - digital (ecl) - digital (lvds) 1 1 4 v cc v plusd v plusd 4.75 5 - 0.8 1.45 5.25 v v v positive supply current - analog - digital 1 1 i vcc i vplusd 138 154 205 200 ma ma negative supply voltage - analog - digital 1 1 v ee d vee -5.25 -5.25 -5 -5 -4.75 -4.75 v v negative supply current - analog - digital 1 1 v ee i dvee 615 160 750 200 ma ma power dissipation - ecl - lvds 1 4p d 4.6 5.0 5.2 5.7 w w analog inputs full-scale input voltage range (differential mode) (0 v common mode voltage) 4 4 v in, v inb - 125 - 125 125 125 mv mv full-scale input voltage range (single-ended input option) (0 v common mode voltage) 4 4 v in, v inb - 250 0 250 mv mv recommended conditions of use (continued) parameter symbol comments min typ max unit
5 ts83102g0b 2101d?bdc?06/04 analog input power level (50 ? single-ended) 4 p in - 2 dbm analog input capacitance (die) 4 c in 0.3 pf input leakage current 4 i in 10 a input resistance - single-ended - differential 4 4 r in r in 49 98 50 100 51 102 ? ? clock inputs logic common mode compatibility for clock inputs differential ecl to lvds clock inputs common voltage range (v clk or v clkb ) (dc coupled clock input) ac coupled for lvds com patibility (common mode 1.2v) 4v cm -1.2 0 0.3 v clock input power level (low-phase noise sinewave input) 50 ? single-ended or 100 ? differential 4p clk -4 0 4 dbm clock input swing (single ended; with clkb = 50 ? to gnd) 4v clk 200 320 500 mv clock input swing (differential voltage) - on each clock input 4 v clk v clkb 141 226 354 mv clock input capacitance (die) 4 c clk 0.3 pf clock input resistance - single-ended - differential ended r clk r clk 45 90 50 100 55 110 ? ? digital inputs (sdaen, pgeb, decb/diode, b/gb, drrb) - logic low - logic high 4 v il v ih -5 -2 -3 0 v v digital inputs (drrb only) logic compatibility negative ecl - logic low - logic high 4 v il v ih -1.810 -1.165 -1.625 -0.880 v v electrical operating char acteristics (continued) v cc = 5v ; v plusd = 0v (unless otherwise specified). adc performances are independent of v plusd common mode voltage and performances are guaranteed within the limits of the specified v plusd range (from -0.9v to 1.7v); v ee = d vee = -5v; v in - v inb = 500 mvpp (full-scale single-ended or differential input); clock inputs differential driven; analog-input single-ended driven. parameter test level symbol min typ max unit
6 ts83102g0b 2101d?bdc?06/04 notes: 1. differential output buffers impedance = 100 ? differential (50 ? single-ended). see figure 46 starting on page 42. 2. histogram testing at fs = 1 gsps, fin = 100 mhz, dnlrms is a component of quantization noise. 3. histogram testing at fs = 50 msps, fin = 25 mhz 4. this range of gain can be set to "1" by using the gain adjust function. digital outputs (1) logic compatibility (depending on v plusd value) differential ecl (v plusd = -0.8v typical) output levels 50 ? transmission lines, 100 ? (2 x 50 ? ) differentially terminated - logic low - logic high - swing (each single-ended output) - common mode 1 1 1 4 v ol v oh v oh - v ol -0.98 200 -.095 -1.17 -0.94 230 -1.05 -1.10 300 -1.15 v v mv v logic compatibility (depending on v plusd value) lvds (v plusd = 1.45v typical) output levels 50 ? transmission lines, 100 ? (2 x 50 ? ) differentially terminated - logic low - logic high - swing (each single-ended output) - common mode max v plusd = 1.525v typ v plusd = 1.45v min v plusd = 1.375v 4 4 4 4 4 4 v ol v oh v oh - v ol 825 200 1190 1125 1090 1310 230 1200 1575 300 1275 1210 mv mv mv mv mv mv dc accuracy dnlrms (2) 4 dnlrms 0.50 0.53 0.55 lsb differential non-linearity (3) 1 dnl+ 1.5 2 lsb integral non-linearity (3) 1inl-- 4.0 - 2.4 lsb integral non-linearity (3) 1inl+ 2.4 4.0 lsb gain central value (4) 1 0.89 0.94 1.1 gain error drift 4 23 35 ppm/c input offset voltage 1 - 10 10 mv electrical operating char acteristics (continued) v cc = 5v ; v plusd = 0v (unless otherwise specified). adc performances are independent of v plusd common mode voltage and performances are guaranteed within the limits of the specified v plusd range (from -0.9v to 1.7v); v ee = d vee = -5v; v in - v inb = 500 mvpp (full-scale single-ended or differential input); clock inputs differential driven; analog-input single-ended driven. parameter test level symbol min typ max unit
7 ts83102g0b 2101d?bdc?06/04 ac electrical characteristics at ambient and hot temperatures (t j max) parameter test level symbol min typ max unit ac analog inputs full power input bandwidth (1) 4 fpbw 3.3 ghz small signal input bandwidth (10% full-scale) (1) 4 ssbw 3.5 ghz gain flatness (2) 4bf 0.2 0.3db input voltage standing wave ratio (3) 4 vswr 1.1 :1 1.2:1 ac performance: nominal condition at ambient and hot temperatures t j max -1 dbfs single-ended in put mode (unless otherwise specified); 50% clock dut y cycle; 0 dbm differential clock (clk, clkb); binar y output data format signal-to-noise and distortion ratio fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4sinad 47 44 43 38 50 48 45 41 db effective number of bits fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4enob 7.5 7.0 6.8 6.1 8.0 7.6 7.1 6.5 bit signal to noise ratio fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4snr 48 45 44 39 50 48 45 41 db
8 ts83102g0b 2101d?bdc?06/04 notes: 1. see ?definition of terms? on page 35. 2. from dc to 1.5 ghz 3. specified from dc up to 2.5 ghz input signal. input vswr is measured on a soldered device. it assumes an external 50 ? 2 ? controlled impedance line, and a 50 ? driving source impedance (s 11 < - 30 db). total harmonic distortion fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4 ithdi 48 48 44 44 54 53 50 49 db spurious free dynamic range fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4 isfdri 50 50 45 45 59 59 53 54 dbc two-tone third-order intermodulation distortion fs = 1.2 gsps fin1 = 995 mhz fin2 = 1005 mhz [-7dbfs] fs = 1.4 gsps fin1 = 745 mhz fin2 = 755 mhz [-7dbfs] fs = 1.4 gsps fin1 = 995 mhz fin2 = 1005 mhz [-7dbfs] fs = 1.4 gsps fin1 = 1244 mhz fin2 = 1255 mhz [-7dbfs] 4imd31 65 65 65 65 dbfs ac electrical characteristics at ambient and hot temperatures (t j max) (continued) parameter test level symbol min typ max unit
9 ts83102g0b 2101d?bdc?06/04 ac performance at cold temperature (t c min) parameter test level symbol min typ max unit ac performance condition -1 dbfs single-ended input mode; 50% clock duty cycle; 0 dbm differential clock (clk, clkb); binary output data format signal-to-noise and distortion ratio fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4sinad 41 40 39 38 43 42 40 39 db effective number of bits fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4enob 6.5 6.3 6.2 6.0 6.8 6.7 6.4 6.2 bit signal to noise ratio fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4snr 45 44 45 43 46 46 46 44 db total harmonic distortion fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4 ithdi 42 41 40 39 44 43 42 41 db spurious free dynamic range fs = 1 gsps fin = 100 mhz fs = 1.4 gsps fin = 700 mhz fs = 1.4 gsps fin = 1950 mhz fs = 2 gsps fin = 2 ghz 4 isfdri 44 43 41 41 46 45 43 43 dbc
10 ts83102g0b 2101d?bdc?06/04 notes: 1. output error amplitude < 6 lsb, fs = 2 gsps, t j = 110c 2. see ?definition of terms? on page 35. 3. 50 ? // c load = 2 pf termination (for each single-ended output). termination load parasitic c apacitance derating value: 50 ps/pf (ecl). see ?timing information? on page 37. 4. tod and tdr propagation times are defined at package input/ outputs. they are given for reference only. see ?propagation time considerations? on page 37. 5. values for td1 and td2 are given for a 2 gsps external clock frequency (50% duty cycle). for different sampling rates, apply the following formula: td1 = t/2 + (|tod - tdr|) and td2 = t/2 + (|tod - tdr|), where t = clock period. this places the ris- ing edge (true/false) of the differential data ready signal in the middle of the output data valid window. this gives maximum setup and hold times for external data acquisition. transient and switching performances parameter test level symbol min typ max unit transient performance bit error rate (1) 4 ber 10 -12 error/ sample adc setting time (v in - v inb = 400 mvpp) 4 ts 1 ns overvoltage recovery time 4 ort 500 ps adc step response rise/fall time (10 - 90%) 80 100 ps overshoot 4% ringback 2% switching performance and characteristics maximum clock frequency (2) f s max 2 2.2 gsps minimum clock frequency (2) 4f s min 150 200 msps minimum clock pulse width (high) 4 tc1 0.2 0.25 2.5 ns minimum clock pulse width (low) 4 tc2 0.2 0.25 2.5 ns aperture delay (2) 4 ta 160 ps aperture uncertainty (2) 4 jitter 150 200 fs rms output rise/fall time for data (20 - 80%) (3) 4 tr/tf 150 200 ps output rise/fall time for data ready (20 - 80%) (3) 4 tr/tf 150 200 ps data output delay (4) 4 tod 360 ps data ready output delay (4) 4 tdr 410 ps 4 itod minus tdri 050100ps output data to data ready propagation delay (5) 4 td1 250 300 350 ps data ready to output data propagation delay (5) 4 td2 150 200 250 ps output data pipeline delay 4 tpd 4.0 clock cycles data ready reset delay 4 trdr 1000 ps
11 ts83102g0b 2101d?bdc?06/04 notes: 1. unless otherwise specified 2. refer to ?ordering information? on page 55. only minimum and maximum values are guaranteed (typic al values are issued from characterization results). table 1. explanation of test levels level explanation 1 100% production tested at 25c (1) (for "c" temperature range) (2) 2 100% production tested at 25c (1) and sample tested at specified temperatures (for "v" temperature ranges (2) ) 3 sample tested only at specified temperatures 4 parameter is guaranteed by design and characterization testing (thermal steady-state conditions at specified temperature) 5 parameter is a typical value guaranteed by design only 6 100% production tested over specified tem perature range (for "b/q" temperature range (2) )
12 ts83102g0b 2101d?bdc?06/04 figure 2. timing diagram note: detailed timing diagrams are provided on page 39. pipeline delay = 4 clock cycles tod td1 td2 tdr gray to binary decoding logic encoding nn+1n+2 nn+1n+2 nn+1n+2 nn+1n+2 nn+1n+2 nn+1n+2 nn+1n+2 nn+1n+2 latch 1 latch 2 latch 3 latch 4 latch 5 latch 6 latch 7 latch 8 data ready outputs internal clock external clock analog input output latches regeneration latches ta n n+1
13 ts83102g0b 2101d?bdc?06/04 table 2. digital coding differential analog input voltage level digital output binary (b/gb = gnd or floating) msb???....lsb out-of-range gray (b/gb = v ee ) msb???....lsb out-of-range > 250.25 mv >top end of full-scale + ? lsb 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 1 250.25 mv 249.75 mv top end of full-scale + ? lsb top end of full-scale - ? lsb 1 1 1 1 1 1 1 1 1 1 0 1 1 1 1 1 1 1 1 1 0 0 1 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 1 0 125.25 mv 124.75 mv 3/4 full-scale + ? lsb 3/4 full-scale - ? lsb 1 1 0 0 0 0 0 0 0 0 0 1 0 1 1 1 1 1 1 1 1 0 1 0 1 0 0 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 0 0 0.25 mv -0.25 mv mid-scale + ? lsb mid-scale - ? lsb 1 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 0 1 1 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 -124.75 mv -124.25 mv 1/4 full-scale + ? lsb 1/4 full-scale - ? lsb 0 1 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 1 1 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 -249.75 mv -250.25 mv bottom end of full-scale + ? lsb bottom end of full-scale - ? lsb 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 0 0 0 < -250.25 mv < bottom end of full-scale - ? lsb 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 0 1 table 3. die mechanical information description data die size 3740 m x 3820 m (15 m) pad size - single pad - double pad 90 m x 90 m 180 m x 90 m die thickness 380 m 25 m back side metallization none metallization - number of layers - material 3 alcu pad metallization alcu passivation oxyde nitride back side potential -5v
14 ts83102g0b 2101d?bdc?06/04 ts83102g0b package description table 4. pin description (cbga 152) symbol pin number function power supplies v cc , v ccth k1, k2, j3, k3, b6, c6, a7, b7, c7, p8, q8, r8 5v analog supply (connected to same power supply plane) gnd b1, c1, d1, g1, m1, q1, b2, c2, d2, e2, f2, g2, n2, p2, q2, a3, b3, d3, e3, f3, g3, n3, p4, q4, r4, a5, p5, q5, p6, q6, p7, q7, r7, b9, b10, b11, r11, p12, a14, b14, c14, g14, k14, p14, q14, r14, b15, q15, b16, q16 analog ground v ee , v eeth h1, j1, l1, h2, j2, l2, m2, c3, h3, l3, m3, p3, q3, r3, a4, b4, c4, b5, c5, a8, b8, c8, c9, p9, q9, c10, q10, r10 -5v analog supply (connected to same power supply plane) v plusd p10, c11, p11, q11, a12, b12, c12, q12, r12, d14, e14, f14, l14, m14, n14 digital positive supply dv ee a13, b13, c13, p13, q13, r13, h14, j14 -5v digital supply analog inputs vin r5 in-phase (+) analog input signal of the differential sample & hold preamplifier vinb r6 inverted phase (-) analog inpu t signal of the differential sample & hold preamplifier clock inputs clk e1 in-phase (+) clock input clkb f1 inverted phase (-) clock input digital outputs d0, d1, d2, d3, d4, d5, d6, d7, d8, d9 d16, e16, f16, g16, j16, k16, l16, m16, n16, p16 in-phase (+) digital outputs d0 is the lsb, d7 is the msb d0b, d1b, d2b, d3b, d4b, d5b, d6b, d7b, d8b, d9b d15, e15, f15, g15, j15, k15, l15, m15, n15, p15 inverted phase (-) digital outputs or c16 in-phase (+) out-of-range output orb c15 inverted phase (-) out-of-range output dr h16 in-phase (+) data ready signal output drb h15 inverted phase (-) data ready signal output additional functions b/gb a11 binary or gray select output format control - binary output format if b/gb is floating or connected to gnd - gray output format if b/gb is connected to v ee
15 ts83102g0b 2101d?bdc?06/04 symbol pin number function decb/diode a10 decimation function enable or die junction temperature measurement: - decimation active when connected to v ee (die junction temperature monitoring is not possible) - normal mode when connected to ground or left floating - die junction temperature monitoring when current is applied pgeb a9 active low pattern generator enable - digitized input delivered at outputs according to b/gb if pgeb is floating or connected to gnd - checker board pattern delivered at outputs if pgeb is connected to v ee drrb n1 asynchronous data ready reset function (active at ecl low level) or when connected to v ee ga r9 gain adjust sda a6 sampling delay adjust sdaen p1 sampling delay adjust enable - inactive if floating or connected to gnd - active if connected to v ee table 4. pin description (cbga 152) (continued)
16 ts83102g0b 2101d?bdc?06/04 figure 3. pinout notes: 1. to simplify pcb routing, the 4 nc balls can be electrically connected to the gnd balls. 2. the pinout is shown from the bottom. the colu mns and rows are defined differently from the jedec standard. ts83102g0bm ci-cga 152 bottom view decb/ diode pgeb orb or
17 ts83102g0b 2101d?bdc?06/04 thermal and moistu re characteristics dissipation by conduction and convection the thermal resistance from junction to ambient rth ja is around 30 c/w. therefore, to lower rth ja , it is mandatory to use an external heat sink to improve dissipation by convection and conduction. the heat sink should be fixed in contact with the top side of the package (cuw heat spreader over al2o3) which is at -5v. the heat sink needs to be electrically isolat ed, using adequate low rth electrical isolation. example: the thermal resistance from case to ambient rth ca is typically 4.0 c/w (0 m/s air flow or still air) with the heat sink depi cted in figure 4 on page 18, of dimensions 50 mm x 50 mm x 22 mm (respectively l x l x h). the global junction to ambient thermal resistance rth ja is: 4.35 c/w rth jc + 2.0 c/w thermal grease resistance + 4.0 c/w rth ca (case to ambi- ent) = 10.35 c/w total (rth ja ). assuming: a typical thermal resistance from the junction to the bottom of the case rth jc of 4.35 c/w (finite element method thermal simulati on results): this value does not include the thermal contact resistance between the pac kage and the external heat sink (glue, paste, or thermal foil interface, for example). as an example, use a 2.0 c/w value for a 50 m thickness of thermal grease. note: example of the calculation of the ambient temperature t a max to ensure t j max = 110 c: assuming rth ja = 10.35 c/w and power dissipation = 4.6 w, t a max = t j - (rth ja x 4.6 w) = 110 - (10.35 x 4.6) = 62.39 c. t a max can be increased by lowering rth ja with an adequate air flow ( 2 m/s, for example).
18 ts83102g0b 2101d?bdc?06/04 figure 4. black anodized aluminium heat sink glued on a copper base screwed on board (all dimensions in mm) note: the cooling system efficiency can be m onitored using the temperature sensing diodes, integrated in the device. refer to ?decb/diode: junction temperature monitoring and output decimation enable? on page 45. thermal dissipation by conduction only when the external heat sink cannot be used, the relevant thermal resistance is the thermal resistance from the junction to the bottom of the balls: rth j-bottom-of-balls . the thermal path, in this case, is the junction, then the silicon, glue, cu w heat spreader, pack- age al2o3, and the balls (sn63pb37). the finite element method (fem) with the thermal simulator leads to rth j-bottom of balls = 12.3c/w. this value assumes pure conduction from the junction to the bottom of the balls (this is the worst case, no radiation and no convection is applied). with such an assumption, rth j- bottom-of-balls is user-independent. to complete the thermal analysis, you must add the thermal resistance from the top of the board (on which the device is soldered) to the ambient resistance, whose values are user- dependent (the type of board, thermal, routing, area covered by copper in each board layer, thickness, airflow or cold plate are all parameters to consider). 8.5 50 52 20 22 15 9 40 0.5 7.4 circular base (diam. 8.5 mm) cuw heat spreader tied to v ee = -5 v ai203 copper base with standoffs board holes for screw (diam. 2 mm) black anodized aluminium
19 ts83102g0b 2101d?bdc?06/04 typical characterization results nominal conditions v cc = 5v; 50% clock duty cycle; binary output data format; t j = 80c; -1 dbfs, unless other- wise specified. typical full power input bandwidth vin = -1 dbfs gain flatness at 0.15 db from dc to 1.5 ghz full power input bandwidth at -3 db > 3.3 ghz figure 5. full power input bandwidth at -3 db typical vswr versus input frequency figure 6. vswr curve for vin and clk -6.0 -5.5 -5.0 -4.5 -4.0 -3.5 -3.0 -2.5 -2.0 -1.5 -1.0 -0.5 0.0 100 300 500 700 900 1100 1300 1500 1 700 1 900 2100 2300 2500 2 700 2 900 3 100 3300 3500 fin (mhz) dbf s gain flatness (0.15 db) -3 db bandwidth 1.0 1.1 1.2 1.3 1.4 1.5 1.6 1.7 0 500 1000 1500 2000 2500 3000 3500 frequency (mhz) vswr clk vin
20 ts83102g0b 2101d?bdc?06/04 typical step response tr measured = 90 ps = sqrt (tr pulsegenerator 2 +tr adc 2 ) tr pulsegenerator = 41 ps (estimated) actual tr adc = 80 ps figure 7. step response (random interleaved sampling method measure) figure 8. zoom on rise time step response note: overshoot and ringback are not measurable (estimated by simulation at 4% and 2% respectively). 0 200 400 600 800 1000 4.00e-15 2.00e-10 4.00e-10 6.00e-10 8.00e-10 1.00e-09 1.20e-09 time (s) lsb 200 300 400 500 600 700 800 4.00e-10 5.00e-10 6.00e-10 7.00e-10 8.00e-10 9.00e-10 1.00e-09 time (s) lsb +90 % +10 % tr adc = 80 ps
21 ts83102g0b 2101d?bdc?06/04 typical dynamic performances versus sampling frequency figure 9. enob versus sampling frequency in nyquist conditions (fin = fs/2) figure 10. sfdr versus sampling frequency in nyquist conditions (fin = fs/2) figure 11. thd versus sampling frequency in nyquist conditions (fin = fs/2) figure 12. snr versus sampling frequency in nyquist conditions (fin = fs/2) 0 1 2 3 4 5 6 7 8 9 400 600 800 1000 1200 1400 1600 1800 2000 fs (msps) enob (bits) -70 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 400 600 800 1000 1200 1400 1600 1800 2000 fs (msps) sfdr (dbc) -70 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 400 600 800 1000 1200 1400 1600 1800 2000 fs (msps) thd (db) 20 25 30 35 40 45 50 55 60 400 600 800 1000 1200 1400 1600 1800 2000 f s (msps) snr (db)
22 ts83102g0b 2101d?bdc?06/04 typical dynamic performances versus fin figure 13. enob versus input frequency at fs = 1.4 gsps and fs = 1.7 gsps figure 14. thd versus input frequency at fs = 1.4 gsps and fs = 1.7 gsps figure 15. sfdr versus input frequency at fs = 1.4 gsps and fs = 1.7 gsps figure 16. snr versus input frequency at fs = 1.4 gsps and fs = 1.7 gsps 2 3 4 5 6 7 8 9 0 200 400 600 800 1000 1200 1400 1600 1800 2000 fin (mhz) enob (bits) fs = 1.4 gsps fs = 1.7 gsps -70 -65 -60 -55 -50 -45 -40 0 200 400 600 800 1000 1200 1400 1600 1800 2000 fin (mhz) thd (db) fs = 1.4 gsps fs = 1.7 gsps -80 -75 -70 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 0 200 400 600 800 1000 1200 1400 1600 1800 2000 fin (mhz) sfdr (dbc) fs = 1.4 gsps fs = 1.7 gsps 30 35 40 45 50 55 60 0 200 400 600 800 1000 1200 1400 1600 1800 2000 fin (mhz) snr (db) fs = 1.4 gsps fs = 1.7 gsps
23 ts83102g0b 2101d?bdc?06/04 typical reconstructed signals and signal spectrum the adc input signal is sampled at a full sampli ng rate, but the output data is 8 or 16 times decimated so as to relax the acquisition syst em data rate. as a c onsequence, the calculation software sees an effective frequency divided by 8 or 16, compared to the adc clock frequency used (fs). the spectrum is thus displayed from dc to fs/2 divided by the decimation factor. decimation only folds all spectral components between dc and fs/2 divided by the decima- tion factor but does not change their amplitude. this does not have any impact on the fft spectr al characteristics because of the ergodicity of the samples (time average = statistic average). the input frequency is chosen to respect the coherence of the acquisition. figure 17. fs = 1.4 gsps and fin = 702 mhz, -1 dbfs; decimation factor = 16, 32 kpoints fft figure 18. fs = 1.4 gsps and fin = 1399 mhz, -1 dbfs; decimation factor = 16, 32 kpoints fft
24 ts83102g0b 2101d?bdc?06/04 figure 19. fs = 1.7 gsps and fin = 898 mhz, -1 dbfs; decimation factor = 16, 32 kpoints fft figure 20. fs = 1.7 gsps and fin = 1699 mhz, -1 dbfs; decimation factor = 8, 32 kpoints fft figure 21. fs = 2 gsps and fin = 1998 mhz, -1 dbfs; decimation factor = 8, 32 kpoints fft
25 ts83102g0b 2101d?bdc?06/04 sfdr performance with/without external dither figure 22. sfdr (in dbc) with and without dither (-23 dbm dc to 5 mhz out of band dither) fs = 1.4 gsps and fin = 710 mhz an increase in sfdr up to >10 db with an addition of -23 dbrms dc to 5 mhz out-of-band dither is noted. the dither profile has to be defined according to the adc?s inl pattern as well as the trade-off to be reached between the increase in sfdr and the loss in snr. please refer to the application note on dither for more information on adding dither to an adc. typical dual tone dynamic performance figure 23. dual tone reconstructed signal spectrum at fs = 1.2 gsps, fin1 = 995 mhz, fin2 = 1005 mhz (-7 dbfs), imd3 = 64 dbfs note: the output data is not dec imated. the spectrum is displayed from dc to 600 mhz. -120 -100 -80 -60 -40 -20 0 0 50 100 150 200 250 300 350 400 450 500 550 600 fs (mhz) dbfs f2 = fs - fin2 = 195 mhz -7 dbfs f1 = fs - fin1 =205 mhz -7 dbfs f1 - f2 10 mhz -75 dbfs 2f2 - f1 185 mhz -64 dbfs 2f1 - f2 215 mhz -65 dbfs f1 + f2 400 mhz -73 dbfs 2f2 + f1 595 mhz -63 dbfs f s/ 2 imd3
26 ts83102g0b 2101d?bdc?06/04 figure 24. dual tone reconstructed signal spectrum at fs = 1.4 gsps, fin1 = 745 mhz, fin2 = 755 mhz (-7 dbfs), imd3 = 65 dbfs note: the adc input signal is sampled at 1.4 gsps but the data acquisition is 8 times decimated. thus, the spectrum is displayed from dc to fs/2 divided by the decimation factor [(fs/2)/8 = 87.5 mhz]. figure 25. dual tone reconstructed signal spectrum at fs = 1.4 gsps, fin1 = 995 mhz, fin2 = 1005 mhz (-7 dbfs), imd3 = 64 dbfs note: the adc input signal is sampled at 1.4 gsps but the data acquisition is 8 times decimated. thus, the spectrum is displayed from dc to fs/2 divided by the decimation factor [(fs/2)/8 = 87.5 mhz]. -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 0 5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 dbfs 2f1 - f2 = 35 mhz -68 dbfs 2f2 - f1 = 65 mh z -65 dbfs f1 - f2 = 10 mhz -78 dbfs f1 + f2 = 75 mhz -68 dbfs 2f2 + f1 = 20 mhz -72 dbfs f2 = - 4 x (fs/8) + fin2 = 55 mhz -7 dbfs f1 = -4 x (fs/8) + fin1 = 45 mhz -7 dbfs mhz 87.5 = fs/16 imd3 -120 -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 0 5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 fs/8 (mhz) dbfs 2f1 - f2 = 65 mhz -65 db 2f2 - f1 = 35 mhz -64 dbfs f1 - f2 = 10 mhz -70 dbfs f1 + f2 = 75 mhz -62 dbfs 2f2 + f1 = 20 mhz -70 dbfs 87.5 = fs/16 f1 = 6 x (fs/8) - fin1 = 55 mhz -7 dbfs f2 = 6 x (fs/8) - fin2 = 45 mhz -7 dbfs imd3
27 ts83102g0b 2101d?bdc?06/04 figure 26. dual tone reconstructed signal spectrum at fs = 1.4 gsps, fin1 = 1244 mhz, fin2 = 1255 mhz (-7 dbfs), imd3 = 65 dbfs note: the adc input signal is sampled at 1.4 gsps but data acquisition is 8 times decimated. thus, the spectrum is displayed from dc to fs/2 divided by the decimation factor [(fs/2)/8 = 87.5 mhz]. the dual tone imd3 at 1.4 gsps is around -65 dbfs for fin = 1 ghz 250 mhz (fin range is from 750 mhz to 1250 mhz). -110 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 10 0 5 10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 mhz dbfs f2 = - 7 x (fs/8) + fin2 = 30 mhz -7dbfs f1 = -7 x (fs/8) + fin = 19 mhz -7dbfs 2f2 + f1 = 79 mhz -60 dbfs 2f2 - f1 = 41 mhz -65 dbfs 2f1 - f2 = 8 mhz -68 dbfs f1 + f2 = 49 mhz -68 dbfs f1 - f2 = 11 mhz -66 dbfs 87.5 = fs/16 2f1 + f2 = 68 mhz -62 dbfs imd3
28 ts83102g0b 2101d?bdc?06/04 typical performance sensitivity ve rsus power supply and temperature figure 27. enob versus junction temperature (fs = 1.4 gsps, fin = 698 mhz, -1 dbfs) figure 28. sfdr versus junction temperature (fs = 1.4 gsps, fin = 698 mhz, -1 dbfs) figure 29. snr versus junction temperature (fs = 1.4 gsps, fin = 698 mhz, -1 dbfs) 3 3.5 4 4.5 5 5.5 6 6.5 7 7.5 8 10 20 30 40 50 60 70 80 90 100 110 tj (?c) bits -70 -60 -50 -40 -30 -20 -10 0 10 20 30 40 50 60 70 80 90 100 110 tj (?c) dbc 30 35 40 45 50 55 60 10 20 30 40 50 60 70 80 90 100 110 tj (?c) db
29 ts83102g0b 2101d?bdc?06/04 figure 30. enob versus v cc and v ee ; fs = 1.4 gsps versus fin (v cc = iv ee i = 4.75v, 5v and 5.25v) figure 31. sfdr versus v cc and v ee ; fs = 1.4 gsps versus fin (v cc = iv ee i = 4.75v, 5v and 5.25v) figure 32. snr versus v cc and v ee ; fs = 1.4 gsps versus fin (v cc = |v ee | = 4.75v, 5v and 5.25v) 8.00 7.50 7.00 6.50 6.00 enob (bits) 5 v 5.25 v 4.75 v fin (mhz) -40.00 -45.00 -50.00 -55.00 -60.00 -65.00 -70.00 sfdr (dbc) fin (mhz) 5 v 5.25 v 4.75 v snr (db) fin (mhz) 5 v 5.25 v 4.75 v
30 ts83102g0b 2101d?bdc?06/04 considerations on enob: linearity and noise contribution figure 33. example of a 16-kpoint fft computatio n at fs = 1.4 gsps, fin = 702 mhz, -1dbfs, t j = 80c; bin spacing = (fs/2) / 16384 = 2.67 khz this is a 16384 points fft. it is 16 times decimated since a demux 1:8 is used to relax the acquisition system data rate, and data is captured on the rising edge of the data ready signal. the spectrum is computed over the first nyquis t zone from dc to fs/2 divided by the decima- tion factor, which equals fs/32 = 43.75 mhz. legend: 1. ideal 10-bit quantization noise spec tral density, peak value = -84 db 2. average snr noise floor: 47 db + 10 log (n fftpoint /2) = 86 db including thermal noise 3. average snr noise floor: 57 db + 10 log (n fftpoint /2) = 96 db without thermal noise 4. ideal 10-bit averaged snr noise floor 6.02 x (n = 10) + 1.76 + 10 log (n fftpoint /2) = 101 db note: the thermal noise floor is expressed in dbm/hz (at t = 300 k, b = 1 hz): 10 log (ktb/1 mw) = -174 dbm/hz or -1 39.75 dbm/2.67 khz. thd is ca lculated over the 25 first harmonics. with adc input referred thermal noise: enob = 7.6 bits sinad = 47 db thd = -55.7 db (over 25 harmonics) sfdr = -62.6 dbc snr = 47.3 db fin = -8 x (fs/16) + 702 mhz = 2 mhz sfdr = -63 dbc 1 2 3 4
31 ts83102g0b 2101d?bdc?06/04 without adc input referred thermal noise: enob = 9.2 bits sinad = 57 db thd = -55.7 db (over 25 harmonics) sfdr = -62.6 dbc snr = 57.3 db conclusion: though the enob is 7.6 bits (in this example at 1.4 gsps nyquist conditions), the adc fea- tures a 10-bit linearity regarding the 60 db typical sfdr performance. however, it has to be pointed out that the enob is actually limited by the adc?s input referred thermal noise, which dominates the rms quantization noise. for certain applications (using a spread spectrum) the signal may be recovered below the thermal noise floor (by cross correla- tion since it is white noise). therefore, the thermal noise can be extracted from the enob: the enob without a referred input thermal noise is 9.2 instead of 7.6 in this example, only limited by the quantization noise and clock induced jitter.
32 ts83102g0b 2101d?bdc?06/04 equivalent input/output schematics figure 34. equivalent analog input circuit and esd protections note: 100 ? termination midpoint is located inside the package cavity and is dc coupled to ground. figure 35. equivalent clock input circuit and esd protections note: 100 ? termination midpoint is on-chip and ac coupled to ground through a 40 pf capacitor. 1 ma 1 ma vee = -5v die pads vin vinb termination resistors soldered into the package cavity 50 ? 2% gnd 50 ? 2% package pins esd 120 ff vee = -5v 1.5v vee = -5v esd 120 ff 50 ? controlled transmission line (bonding + package + ball) double pad 260 ff 50 ? controlled transmission line (bonding + package + ball) double pad 260 ff 50 ? 50 ? vee = -5v 150 ? 150 ? vee = -5v 40 pf 400 a 400 a vee = -5v double pad 260 ff esd 120 ff esd 215 ff esd 120 ff clk clkb double pad 260 ff double pad 260 ff mid vee = - 5v
33 ts83102g0b 2101d?bdc?06/04 figure 36. equivalent data output buffer circuit and esd protections figure 37. adc gain adjust equivalent input circuits and protections 10.5 ma outb 50 ? 50 ? vplusd vplusd vplusd out pad 130 ff pad 130 ff esd 100 ff esd 100 ff esd 60 ff dvee = -5v esd 60 ff -+ vcc = 5v vee = -5v 1 k ? 20 ? ga vee = -5v pad 130 ff esd 75 ff esd 65 ff 0.9v gnd 0v 100 a 10 pf 100 a
34 ts83102g0b 2101d?bdc?06/04 figure 38. b/gb and pgeb equivalent input schematics and esd protections figure 39. drrb equivalent input schema tics and esd protections 5 k ? 1 k ? 2 k ? gnd gnd gnd b/gb vee = -5v vee = -5v esd 75 ff esd 65 ff pad 130 ff 250 a 250 ? -1.3v 200 ? 10 k ? gnd drrb gnd gnd vee = -5v vee = -5v -1.3v -2.6v 200 a 200 ? pad 130 ff esd 65 ff esd 75 ff
35 ts83102g0b 2101d?bdc?06/04 definition of terms table 5. definitions of terms term description ber bit error rate probability to exceed a specified error threshold for a sample. an error code is a code that differs by more than 4 lsb from the correct code bw full-power input bandwidth the analog input frequency at which the fu ndamental component in the digitally reconstructed output has fallen by 3 db with respect to its low frequency value (determined by fft analysis) for input at full-scale dg differential gain the peak gain variation (in percent) at five different dc levels for an ac signal of 20% full- scale peak to peak amplitude. f in = 5 mhz (tbc) dnl differential non- linearity the differential non-linearity for an output c ode (i) is the difference between the measured step size of code (i) and the ideal lsb step size. dnl (i) is expressed in lsbs. dnl is the maximum value of all dnl (i). dnl error specif ication of less than 1 lsb guarantees that there are no missing output codes and th at the transfer function is monotonic dp differential phase the peak phase variation (in degrees) at five different dc levels for an ac signal of 20% full- scale peak to peak amplitude. f in = 5 mhz (tbc) fs max maximum sampling frequency sampling frequency for which enob < 6 bits fs min minimum sampling frequency sampling frequency for which the adc gain has fallen by 0.5 db with respect to the gain reference value. performances are not guaranteed below this frequency fpbw full power input bandwidth analog input frequency at which the fundamental component in the digitally reconstructed output waveform has fallen by 3 db with respect to its low frequency value (determined by fft analysis) for input at full-scale -1 db (-1 dbfs) enob effective number of bits where a is the actual input amplitude and v is the full-scale range of the adc under test imd3 inter modulation distortion the two tones third order intermodulation distor tion (imd3) rejection is the ratio of either input tone to the worst third order intermodulation products inl integral non-linearity the integral non-linearity for an output code (i ) is the difference between the measured input voltage at which the transition occurs a nd the ideal value of this transition. inl (i) is expressed in lsbs, and is the maximum value of all inl (i) jitter aperture uncertainty the sample to sample variation in aperture delay. the voltage error due to jitter depends on the slew rate of the signal at the sampling point npr noise power ratio the npr is measured to characterize the adc?s performance in response to broad bandwidth signals. when using a notch-filt ered broadband white-noise generator as the input to the adc under test, the noise-to-power ratio is defined as the ratio of the average out-of-notch to the average in-notch power spectral density magnitudes for the fft spectrum of the adc output sample test nrz non return to zero when the input signal is larger than the upper bound of the adc input range, the output code is identical to the maximum code and the out-of-range bit is set to logic one. when the input signal is smaller than the lower bound of the adc input range, the output code is identical to the minimum code, and the out-of-range bit is se t to logic one (it is assumed that the input signal amplitude remains within the absolute maximum ratings) ort overvoltage recovery time time to recover 0.2% accuracy at the output, after a 150% full-scale step applied on the input is reduced to midscale e nob sinad 1.76 ? 20 a fs 2 ? ---------------- log + 6.02 -------------------------------------------------------------------------- - =
36 ts83102g0b 2101d?bdc?06/04 psrr power supply rejection ratio psrr is the ratio of input offset variat ion to a change in power supply voltage sfdr spurious free dynamic range the ratio expressed in db of the rms signal amplitude, set at 1 db below full-scale, to the rms value of the next highest spectral component (peak spurious spectral component). sfdr is the key parameter for selecting a co nverter to be used in a frequency domain application (radar systems, digital receiver, network analyzer...). it may be reported in dbc (i.e., degrades as signal level is lowered), or in dbfs (i.e. always related back to converter full-scale) sinad signal to noise and distortion ratio the ratio expressed in db of the rms signal amplitude, set to 1 db below full-scale, to the rms sum of all other spectral components, including the harmonics except dc snr signal to noise ratio the ratio expressed in db of the rms signal amplitude, set to 1 db below full-scale, to the rms sum of all other spectral compone nts excluding the first five harmonics ssbw small signal input bandwidth analog input frequency at which the fundamental component in the digitally reconstructed output waveform has fallen by 3 db with respect to its low frequency value (determined by fft analysis) for input at full-scale -10 db (-10 dbfs) ta aperture delay the delay between the rising edge of the differential clock inputs (clk,clkb) (zero crossing point), and the time at which (v in , v inb ) is sampled tc encoding clock period tc1 = minimum clock pulse width (high) tc = tc1 + tc2 tc2 = minimum clock pulse width (low) td1 time delay from data to data ready general expression is td1 = tc1 + tdr - tod with tc = tc1 + tc2 = 1 encoding clock period td2 time delay from data ready to data general expression is td1 = tc1 + tdr - tod with tc = tc1 + tc2 = 1 encoding clock period tf fall time time delay for the output data signals to fall from 80% to 20% of delta between low level and high level thd total harmonic distortion the ratio expressed in dbc of the rms sum of the first five harmonic components, to the rms value of the measured fundamental spectral component tod digital data output delay the delay from the falling edge of the differential clock inputs (clk, clkb) (zero crossing point) to the next point of change in the differential output data (zero crossing) with a specified load tpd pipeline delay the number of clock cycles between the samplin g edge of an input data and the associated output data being made available (not taking in account the tod). for the jts8388b the tpd is 4 clock periods tr rise time time delay for the output data signals to rise from 20% to 80% of delta between the low level and high level trdr data ready reset delay delay between the falling edge of the data ready output asynchronous reset signal (ddrb) and the reset to digital zero transition of the data ready output signal (dr) ts settling time time delay to achieve 0.2% accuracy at the converter output when an 80% full-scale step function is applied to the differential analog input vswr voltage standing wave where s11 is the reflection coefficient of the scattering matrix. the vswr over frequency measures the degree of mismatching between the packaged adc input impedance (ideally 50 ? or so) and the transmission line?s impedance. the packaged adc input impedance (transmission line and termination) is controlled so as to ensure vswr < 1.2 :1 from dc up to 2.5 ghz. a vswr of 1.2 :1 corresponds to a 0.039 db insertion loss (20 db return loss) - i.e. 99% power transmitted and 1% reflected table 5. definitions of terms (continued) vswr 1 s 11 + () 1 s 11 ? () =
37 ts83102g0b 2101d?bdc?06/04 ts83102g0b operating features timing information timing value for ts83102g0b the timing values are defined in the ?electrical operating characteristics? on page 4. the timing values are given at the package i nputs/outputs, taking into account the package?s transmission line, bond wire, pad and esd protections capacitance, as well as specified termi- nation loads. the evaluation board propagation delays in 50 ? controlled impedance traces are not taken into account. you should apply proper derating values corresponding to termina- tion topology. propagation time considerations the tod and tdr timing values are given from the package pin to pin and do not include the additional propagation times between the device pi ns and input/output termination loads. for the evaluation board, the propagation time delay is 6.1 ps/mm (155 ps/inch) corresponding to a 3.4 dielectric constant (at 10 ghz) of the ro4003 used for the board. if a different dielectric layer is used (for instance teflon), you should use appropriate propaga- tion time values. td1 and td2 do not depend on propagation times because they are differential data (see ?definition of terms? on page 35). td1 and td2 are also the most straightforward data to measure, because they are differential: td can be measured di rectly on the terminat ion loads, with matching oscilloscope probes. tod-tdr variation over temperature values for tod and tdr track each other over the temperature (there is a 1% variation for tod and tdr per 100 c temperature variation). therefore the tod and tdr variation over temperature is negligible. moreover, the internal (on-chip) skews between each tod and tdr data effect can be considered negligible. consequently, the minimum values for tod and tdr are never more than 100 ps apart. the same is true for their maximum values. however, the external tod and tdr values can be dictated by the total digital data skews between each tod and tdr. these digital skews can include the mcm board, bonding wires and output line length differences, as well as output termination impedance mismatches. the external (on-board) skew effect has not be en taken into account for the specification of tod and tdr minimum and maximum values. principle of operation the analog input is sampled on the rising edge of the external clock?s input (clk/clkb) after ta (aperture delay). the digitized data is avai lable after 4 clock periods? latency (pipeline delay [tpd]) on the clock?s rising edge, afte r a typical propagation delay tod. the data ready differential output signal frequency (dr/drb) is half the external clock?s frequency. it switches at the same rate as the digital outputs. the data ready output signal (dr/drb) switches on the external clock?s fallin g edge after a prop agation delay tdr. if tod equals tdr, the rising edge (true-false) of the differential data ready signal is placed in the middle of the output data valid window . this gives maximum setup and hold times for external data acquisition. a master asynchronous reset input command drrb (ecl compatible single-ended input) is available for initializing the differential data r eady output signal (dr/drb). this feature is mandatory in certain applications using interleaved adcs or using a single adc with demulti- plexed outputs. without data ready signal initiali zation, it is impossible to store the output digital data in a defined order.
38 ts83102g0b 2101d?bdc?06/04 when used with atmel?s ts81102g0 1:4/8 8/10 bit dmux, it is not necessary to initialize data ready, as this device can start on either clock edge. principle of data ready signal control by drrb input command data ready output signal reset the data ready signal is reset on the drrb input command?s falling edge, on the ecl logical low level (-1.8v). drrb may also be tied to v ee = - 5v for the data ready output signal mas- ter reset. as long as drrb remains at a logical low level, (or tied to v ee = - 5v), the data ready output remains at a logical zero and is independent of the external free-running encod- ing clock. the data ready output signal (dr/drb) is reset to a logical zero after trdr. trdr is measured between the -1.3v point of the drrb input command?s falling edge and the zero crossing point of the differential data ready output signal (dr/drb).the data ready reset command may be a pulse of 1 ns minimum time width. data ready output signal restart the data ready output signal restarts on the drrb command?s rising edge, on the ecl logi- cal high level (-0.8v). drrb may also be grounded, or may float, for normal free-running of the data ready output signal. the data ready signal?s restart sequence depends on the logical level of the external encoding clock, at a drrb rising edge instant: the drrb?s rising edge occurs when the ex ternal encoding clock input (clk/clkb) is low : the data ready output?s first rising edge occurs after half a clock period on the clock?s falling edge, and a tdr delay time of 410 ps, as defined above. the drrb?s rising edge occurs when the ex ternal encoding clock input (clk/clkb) is high : the data ready output?s first rising edge occurs after one clock period on the clock?s falling edge, and a tdr delay time of 410 ps. consequently, as the analog input is sampled on the clock?s rising edge, the first digitized data corresponding to the first acquisition (n), afte r a data ready signal restart (rising edge), is always strobed by the third rising edge of the data ready signal. the time delay (td1) is specified between the la st point of a change in the differential output data (zero crossing poin t) to the rising or falling edge of the differential da ta ready signal (dr/drb) [zero crossing point]. note: for normal initialization of the data ready ou tput signal, the external encoding clock signal fre- quency and level must be controlled. the minimum encoding clock sampling rate for the adc is 150 msps, due to the internal sample and hold drop rate. consequently the clock cannot be stopped.
39 ts83102g0b 2101d?bdc?06/04 timing diagram figure 40. ts83102g0b timing diagram (2 gsps clock rate ) - data ready reset clock held at low level figure 41. ts83102g0b timing diagram (2 gsps clock rate ) - data ready reset clock held at high level n - 4 n - 3 n - 2 n - 1 n n + 1 v in /v inb clk/clkb digital outputs data ready dr/drb data ready reset ta = 160 ps n n + 1 n + 2 n + 3 n - 5 tod = 360 ps tdr = 410 ps trdr = 1000 ps 1 ns tc = 500 ps tc1 tc2 tpd = 4.0 clock period tod = 360 ps 500 ps tdr = 410 ps td1 = tc1 + tdr - tod = tc1 + 50 ps = 300 ps td2 = tc2 + tod - tdr = tc2 - 50 ps = 200 ps n - 4 n - 3 n - 2 n - 1 n n + 1 n - 5 500 ps td2 = tc2 + tod - tdr = tc2 - 50 ps = 200 ps td1 = tc1 + tdr - tod = tc1 + 50 ps = 300 ps tdr = 410 ps tpd = 4.0 clock periods tod = 360 ps trdr = 1000 ps 1 ns tdr = 410 ps tod = 360 ps ta = 160 ps n n + 1 n + 2 n + 3 tc = 500 ps tc1 tc2 v in /v inb clk/clkb digital outputs data ready dr/drb data ready reset
40 ts83102g0b 2101d?bdc?06/04 analog inputs (vin/vinb) static issues: differential versus single-ended (full- scale inputs) the adc?s front-end tra ck and hold differential preamplifier has been designed to be entered either in differential or single-ended mode, up to the maximum operating speed of 2.2 gsps, without affecting dynamic performances (it does not require a single to differential balun). in a single-ended input configuration, the in-phase full-scale input amplitude is 0.5v peak-to- peak, centered on 0v (or -2 dbm into 50 ? ). figure 42. typical single-ended analog input configuration (full-scale) the analog full-scale input range is 0.5v peak-to-peak (vpp), or -2 dbm into the 50 ? (100 ? differential) termination resistor. in the differential mode input configuration, this means 0.25v on each input, or 125 mv around 0v. the input common mode is ground. figure 43. differential inputs volt age span (full-scale) dynamic issues: input impedance and vswr the ts83102g0b analog input features a 100 ? (2%) differential input impedance (2 x 50 ? // 0.3 pf). each analog input (vin,vinb) is terminated by 50 ? single-ended (100 ? differential) resistors (2% matching) soldered into the package cavity. the transmission lines of the adc package?s analog inputs feature a 50 ? controlled imped- ance. each single-ended die input pad capacitance (taking into account the esd protection) is 0.3 pf. this leads to a global input vswr (including ball, package and bounding) of less than 1.2 from dc up to 2.5 ghz. 500 mv full-scale analog input +250 mv -250 +250 mv vin vinb = 0v t mv +125 -125 500 mv full-scale analog input t vin vinb +250 mv -250 mv 0v
41 ts83102g0b 2101d?bdc?06/04 clock inputs (clk/clkb) the ts83102g0b clock inputs are designed for either single-ended or differential operation. the device?s clock inputs are on-chip 100 ? (2 x 50 ? ) differentially terminated. the termina- tion mid point is ac coupled to ground through a 40 pf on-chip capacitor. therefore, either ground or different common modes can be used (ecl, lvds). note: as long as v ih remains below the 1v peak, the adc clock can be dc coupled. if v ih is higher than the 1v peak, it is necessary to ac couple the signal via 100 pf capacitors, for example, and to bias clk and clkb: - clk biased to ground via a 10 k ? resistor - clkb biased to ground via a 10 k ? resistor and to v ee via a 100 k ? resistor. however, logic ecl or lvds square wave clock generators are not recommended because of poor jitter performances. furthermore, the propagation times of the biasing tees used to offset the common mode voltage to ecl or lvds levels may not match. a very low-phase noise (low jitter) sinewave input signal should be used for enhanced snr performance, when digitiz- ing high frequency analog inputs. typically, when using a si newave oscillator featuring a -135 dbc/hz phase noise, at 20 khz from the carri er, a global jitter value (including the adc and the generator) of less than 200 fs rms has been measured. if the clock signal frequency is at fixed rates, it is recommended to narrow-band filter the signal to improve jitter performance. note: the clock input buffer?s 100 ? termination load is on-chip and mid-point ac coupled (40 pf) to the chip?s ground plane, whereas the analog input buffer?s 100 ? termination is soldered inside the package cavity and mid-point dc coupled to the package ground plane.therefore, driving the analog input in single-ended mode does not pe rturb the chip?s ground plane (since the ter- mination mid-point is connected to the package ground plane). however, driving the clock input in single-ended mode does perturb the chip?s gr ound plane (since the termination mid-point is ac coupled to the chip?s ground plane). therefore, it is required to drive the clock input in differ- ential mode for minimum chip ground plane perturbation (a 4 dbm maximum operation is recommended). the typical clock input power is 0 dbm. the minimum operating clock input power is -4 dbm (equivalent to a 250 mv minimum swing amplitude), to avoid snr performance degradations linked to the clock signal?s slew rate. a single to differential balun with sqrt (2) ratio may be used (featuring a 50 ? input impedance with 100 ? differential termination). for instance: 4 dbm is equivalent to 1 vpp into 50 ? and 1.4 vpp into 100 ? termination (secondary). 0 dbm is equivalent to 0.632 vpp into 50 ? and 0.632 x sqrt (2) = 0.894 vpp into 100 ? ter- mination (secondary), 0.226v at each clock input. the recommended clock input?s common mode is ground. differential clock inputs voltage levels (0 dbm typical) figure 44. differential clock inputs - ground common mode (recommended) v +0.23 -0.23 clk clkb 0v t
42 ts83102g0b 2101d?bdc?06/04 equivalent single- ended clock input voltage levels (0 dbm typical) figure 45. single-ended clock inputs - ground common mode noise immunity information the circuit?s noise immunity performance begins at the design level. efforts have been made on the design to make the device as insensitiv e as possible to chip environment perturbations, which may result from the circuit itself or be induced by external circuitry (cascode stage?s iso- lation, internal damping resistors, clamps , internal on-chip decoupling capacitors.) furthermore, the fully differential operation from the analog input up to the digital output pro- vides enhanced noise immunity by common mode noise rejection. the common mode noise voltage induced on the differential analog and cl ock inputs is cancell ed out by these balanced differential amplifiers. moreover, proper active signal shielding has been provided on the chip to reduce the amount of coupled noise on the active inputs. the anal og and clock inputs of the ts83102g0b device have been surrounded by ground pins, which must be directly connected to the external ground plane. digital outputs: terminatio n and logic compatibility each single-ended output of the ts83102g0b?s differential output buffers are internally 50 ? terminated, and feature a 100 ? differential output impedance. the 50 ? resistors are con- nected to the vplusd digital power supply. the ts83102g0b output buffers are designed to drive 50 ? controlled impedance lines properly terminated by a 50 ? resistor. a 10.5 ma bias current flowing alternately into one of the 50 ? resistors when switching, ensures a 0.25v single-ended voltage drop across the resistor (0.5v differential). each single-ended output transmission line length must be kept identical (< 3 mm). mis- matches in the differential line lengths may cause variations in the output differential common mode. it is recommended to bypass the midpoint of the differential 100 ? termination with a 47 pf capacitor, so as to avoid common mode perturbations in case of a slight mismatch in the dif- ferential output line lengths. v +0.32 -0.32 clk clkb 0v t
43 ts83102g0b 2101d?bdc?06/04 see the recommended termination scenarios in figures 46. and 47. below. note: since the output buffers feature a 100 ? differential output impedance, it is possible to directly drive high the input impedance storing registers without terminating the 50 ? transmission lines. timewise, this means that the incident wave reflects at the 50 ? transmission line output and travels back to the 50 ? data output buffer. since the buffer output impedance is 50 ? , no back reflection occurs and the output swing is doubled. vplusd digital power supply settings for differential ecl digital output levels: v plusd should be supplied with -0.8v (or connected to ground via a 5 ? resistor to ensure the -0.8 voltage drop). for the lvds digital out put logic compatibility: v plusd should be tied to 1.45v (75 mv). if used with the ts81102g0 dmux, v plusd can be set to ground. ecl differential output termination configurations figure 46. 50 ? terminated differential outputs (recommended) figure 47. unterminated differentia l outputs (optional) 10.5 ma zc = 50 ? out vplusd = - 0.8v zc = 50 ? 50 ? 50 ? outb 50 ? 50 ? vol typ = -1.17v voh typ = -0.94v differential output swing: 0.23v = 0.46 vpp common mode level = -1.05v 47 pf 10.5 ma zc = 50 ? out vplusd = -0.8v zc = 50 ? outb 50 ? ? vol typ = -1.4v voh typ = -0.94v differential output swing: 0.46v = 0.92 vpp common mode level = -1.17v
44 ts83102g0b 2101d?bdc?06/04 lvds differential output loading configurations figure 48. 50 ? terminated differential outputs (recommended) figure 49. unterminated differentia l outputs (optional) lvds logic compatibility figure 50. lvds format (refer to the ieee standar ds 1596.3 - 1994): 1125 mv < common mode <1275 mv and 250 mv < output swing < 400 mv 10.5 ma zc = 50 ? out vplusd = 1.45v zc = 50 ? 50 ? 50 ? outb 50 ? 50 ? vol typ = 1.09v voh typ = 1.31v differential output swing: 0.23 vp = 0.46 vpp common mode level = 1.20v 47 pf 10.5 ma zc=50 ? out vplusd = 1.45v zc=50 ? outb 50 ? ? vol typ = 0.85v voh typ = 1.31v differential output swing: 0.46v = 0.92 vpp common mode level = -1.08v common mode each single-ended output swing max voh max = 1.575v swing min voh min = 1.575v vol max = 1.075v vol min = 0.825v cm max = 1275 mv cm min = 1125 mv output swing max = 300 mvp output swing min = 200 mv p 0v true-false output false-true output swing max
45 ts83102g0b 2101d?bdc?06/04 main functions of the adc out-of-range bit (or/orb) the out-of-range bit reaches a logical high state when the input exceeds the positive full-scale or falls below the negative full-scale. when the analog input exceeds the positive full-scale, the digital outputs remain at a logical high state with or/orb at a logical one. when the ana- log input falls below the negative full-scale, the digital outputs remain at a logical low state, with or/orb at a logical one again. bit error rate (ber) the ts83102g0b?s internal regeneration latches indecisions (for inputs very close to the latches? threshold). this may produce errors in the logic encoding circuitry, leading to large amplitude output errors. this is because the latches regenerate the internal analog residues into logical states with a finite voltage gain value (av) within a given positive amount of time d(t): av = exp (d (t)/t), with t being the positive regeneration time constant feedback. the ts83102g0b has been designed to reduce the probab ility of such errors occuring to 10-12 (measured for the converter at 2 gsps). a standard technique for reducing the ampli- tude of such errors down to 1 lsb consists in setting the digital output data to gray code format. however, the ts83102g0b has been designed to feature a bit error rate of 10-12 with a binary output format. gray or binary output data format selection to reduce the amplitude of such errors when they occur, it is possible to choose between the binary or gray output data format by storing gray output codes. digital data format selection: binary output format if b/gb is floating or gnd. gray output format if b/gb is connected to v ee . pattern generator function the pattern generator function (enabled by connecting pin a9 pgeb to v ee = -5v) allows you to rapidly check the adc?s operation thanks to a checker board pattern delivered internally to the adc. each of the adc?s output bits should toggle from 0 to 1 successively, giving sequences such as 0101010101 and 1010101010 ever y 2 cycles. this function is disabled when pgeb is left floating or connected to ground. decb/diode: junction temperature monitoring and output decimation enable the decb/diode pin is provided to enable the decimation function and monitor the die junc- tion temperature. when v ee = -5v, the adc runs in ?decimation by 32? mode (1 out of 32 data is output from the adc, thus reducing the data rate by 32). when the decb/diode pin is left floating or connected to ground, then the adc is said to be in a "normal" mode of operation (the output data is not decimated) and can be used for die junction temperature monitoring only. if you do not intend to use the die junction temperature monitoring function, the decb/diode pin (a10) has to be left either floating or connected to ground. the decimation function can be used to debug the adc at initial stages. this function enables you to reduce the adc output rate by 32, thus reducing the time of the adc?s debug phase at the maximum speed rate, and is compatible with industrial testing environments.
46 ts83102g0b 2101d?bdc?06/04 when this function is active, the adc outputs only 1 out of 32 bits of data, resulting in a data rate 32 times slower than the clock rate. note: the adc decimation test mode is different fr om the pattern generator f unction, which is used to check the adc?s outputs. external configuration description because of the use of one internal diode-mounted transistor (used for junction temperature monitoring), you have to implement external head-to-tail protection diodes so as to avoid potential reverse current flows, which can damage the internal diode component. two external configurations are possible: configuration 1: allows both junction temperature monitoring and output data decimation. configuration 2: allows junction temperature monitoring only. configuration 1 this external configuration allows you to appl y the requested levels to activate output data decimation (v ee = -5v) and at the same time monitor the junction temperature diode (this explains why 7 protection diodes are needed in the other direction, as shown in figure 51). figure 51. recommended diode pin implementation allowing for both die junction temper- ature monitoring function and decimation mode figure 52. diode pin implementation for decimation activation adc pin 1 ma a10 gnd vgnd v vdiode idiode ignd adc pin a10 gnd vee = -5 v
47 ts83102g0b 2101d?bdc?06/04 configuration 2: note: in the preliminary specification, atmel re commends the use of 2 x 3 head-to-tail protection diodes. figure 53. diode pin implementation of die juncti on temperature monitoring function only junction temperature diode transfer function the forward voltage drop (v diode ), across the diode component, versus the junction tempera- ture (including the chip?s parasitic resistance) is given in the following graph (i diode = 1 ma). figure 54. junction temperature versus diode voltage for l = 1 ma adc pin 1ma a10 gnd vgnd v vdiode idiode ignd 79 0 800 810 820 830 840 850 860 870 880 890 900 910 920 930 940 950 -10 0 10 20 30 40 50 60 70 80 90 100 110 jonction temperature (c) diode voltage (mv)
48 ts83102g0b 2101d?bdc?06/04 adc gain control the adc gain is adjustable by using pin r9 of the cbga pac kage. the gain adjust transfer function is shown below. figure 55. gain adjust transfer function sampling delay adjust the sampling delay adjust (sda pin) enables you to fine-tune the sampling adc aperture delay tad around its nominal value (160 ps). this functionality is enabled with the sdaen signal, which is active when tied to v ee and inactive when tied to gnd. this feature is particularly in teresting for interleav ing adcs to increas e the sampling rate. the variation of the delay around its nominal value as a function of the sda voltage is shown in figure 56 (simulation result). figure 56. typical tuning range (120 ps for applied control voltage varying between -0.5v and 0.5v on the sda pin) 0.50 0.60 0.70 0.80 0.90 1.00 1.10 1.20 1.30 -0.5 -0.4 -0.3 -0.2 -0.1 0 0.1 0.2 0.3 0.4 0.5 v ga gain adjust voltage (v) adc gain min typical 400 p 300 p 200 p 100 p -500 m -400 m -300 m -200 m -100 m 0.00 100 m 200 m 300 m 400 m 500 delay in the variable delay cell at 60 c sda voltage delay(s)
49 ts83102g0b 2101d?bdc?06/04 tsev83102g0b evaluation board figure 57. schematic board view note: for more details, refer to the tsev83102g0bgl evaluation board datasheet. adc 10 bits 2 gsps packaged evaluation board general design without drivers gnd 6 66.30 mm . differential clock inputs clkb 50.80 mm . 37.60 mm . 17.40 mm board size : 12.0 x 15.0 cm 4 holes on 44.0 mm square, diam 2.2 for heatsink mounting / centered on packaged device 78.00 mm cal1 drrb 50.00 mm 34.50 mm 17.40 mm control line cal2 differential analog inputs 61.60 mm vin vinb 50 ohm microstrip lines gnd offset adjust length 50 +/- 0.2 mm same length +/- 0.2mm vin single 42.0 mm length 10.00 mm 5.00 mm . 5.00 mm . clk gnd vee 3.00 mm 3.00 mm adjust sampling delay 50 ohm microstrip lines gnd 42.0 +/- 0.2mm same length = cibel 2000.xx test sda gnd component side copyright made in france thomson/tcs 2gsps adc 2000-xx-a 1 dvee package axe gnd vcc vplusd 150.00 mm differential data outputs including data ready gnd 54.00mm 25.00mm adc gain adjust 50 ohm termination resistor 50 ohm microstrip lines gnd gnd 48 48 d5 gnd d6 gnd gnd d6b d7 db7 gnd gnd gnd gnd d9 gnd d9b d8 d8b gnd d5b gnd d4 d4b gnd gnd gnd gnd gnd gnd gnd gnd 7 2 mm bou t de piste vcc b/gb diode v-gnd test gnd 120.0 mm v-diode i-diode b/gb i-gnd gnd veet gnd 71.0 mm 48 gnd gnd d2 d3 gnd d3b drb gnd gnd dr gnd d1b gnd d1 d2b d0 gnd gnd d0b gnd gnd pc pcb gnd gnd gnd gnd 1 gnd gnd 2 x 48 pins connector 2.54 mm pitch vdd 2 mm banana 66 m m + /- 5 m m 66 mm + /- 5 mm 66 mm +/- 5 mm gain sda
50 ts83102g0b 2101d?bdc?06/04 applying the ts83102g0b with the ts81102g0 demultiplexer the ts83102g0b output data rate can be demultiplexed 4 or 8 times by using the ts81102g0 (8/10-bit parallel channel 2 gsps 1:4/1:8 demultiplexer). the adc?s evaluation of static and dynamic performances can be done using the tsev83102g0bgl adc evaluation board, coup led with the ts81102g 0 dmux evaluation board and an acquisition system. the following block diagram shows a typical characterization set-up. figure 58. characterization setup a separate technical specification of the ts81102g0 demultiplexer is available. refer to this document for further information on the device. note: for more information, refer to the ?demux and adcs application notes?. adc clk synchronization data ready data out data out vin data in adc board demux board high speed acquisition system hf oscillo 1 ghz clkin 2 ghz 8
51 ts83102g0b 2101d?bdc?06/04 package description hermetic cbga 152 outline dimensions figure 59. mechanical description bottom view ceramic body size : 21 x 21 mm ball pitch : 1.27 mm cofired : al2o3 optional: discrete capacitor mounting lands on the top side of the package for extra decoupling. metalic cap 9.27 x 9.27 mm chamfer 0.4 (x 4) 1 pin a1 index (no ball) a 21.00 mm 0.20 152 x o d = 0.80 0.10 mm 0.20 0.15 t t a b (position of array of columns/ref a and b) (position of balls within array) 1.27 mm pitch 21.00 mm 0.20 - b - - a -
52 ts83102g0b 2101d?bdc?06/04 figure 60. isometric view figure 61. package top view 4.335 mm marking area 1 21.00 mm sq 4.335 mm marking area 2 6.815 mm cuw 7.2 mm sq is brazed on 9.0 mm sq metallization cuw is connected to v ee pin a1 index (0.50 mm full circle) 9.270 mm 9.085 mm 9.00 mm sq 7.20 mm sq 10.685 mm 2.50 mm 5.605 mm these lands are designed for discrete capacitor device 0603 size (1.6 x 0.8 mm) 2.50 mm 2.50 mm 2.50 mm
53 ts83102g0b 2101d?bdc?06/04 figure 62. package top view with optional discrete capacitors note: for additional decoupling of power supplies, ex tra land capacitors can be used, as shown in fig- ure 62. they are not required if following the ev aluation board?s decoupling recommendations or if using standard power supply sources (performan ce results of the device have proven to be equivalent without these capacitors). marking area 2 marking area 1 4.335 mm 21.00 mm sq 7.20 mm sq 9.00 mm sq 2.50 mm 6.815 mm 9.270 mm 2.50 mm 10.685 mm 4.335 mm 2.50 mm 9.085 mm 2.50 mm cuw 7.2 mm sq is brazed on 9.0 mm sq metalization cuw is connected to v ee 5.605 mm capacitor discrete devices are 0603 size (1.6 x 0.8 mm) thickness 0.8 mm weight 3 - 4 mg each pin a1 index (0.50 mm full circle)
54 ts83102g0b 2101d?bdc?06/04 figure 63. cross section cbga 152 21x21 mm cross section 10 bits/2 gsps adc. external heatsink required low t? solder balls diam 0.76 mm on 1.27 mm grid combo lid soldered 9.27 mm sq 0.254 mm thick grounded al2o3 ceramic cuw heat spreader brazed on al2o3 at vee=-5 volt potential location for external heatsink 1.25 +/- 0.12 mm 0.65 mm 0.50 +/- 0.05 mm 1 .2 7 mm 0. 80 mm 0. 25 0.15
55 ts83102g0b 2101d?bdc?06/04 ordering information part number package temperature range screening level comments ts83102g0bcgl cbga 152 ?c? 0c JTS83102G0-1V1B die ambient visual inspection upon request only (please contact your local atmel sales office)
printed on recycled paper. disclaimer: atmel corporation makes no warranty for the use of its products , other than those expressly contained in the company?s standar d warranty which is detailed in atmel?s terms and conditions locat ed on the company?s web site. the company assumes no responsibi lity for any errors which may appear in this document, reserves the right to change devices or specifications detailed herein at any time wi thout notice, and does not make any commitment to update the information contained her ein. no licenses to patents or other intellectual property of atmel are granted by the company in connection with the sale of atmel produc ts, expressly or by implication. atmel?s products are not aut horized for use as critical components in life support devices or systems. atmel corporation atmel operations 2325 orchard parkway san jose, ca 95131, usa tel: 1(408) 441-0311 fax: 1(408) 487-2600 regional headquarters europe atmel sarl route des arsenaux 41 case postale 80 ch-1705 fribourg switzerland tel: (41) 26-426-5555 fax: (41) 26-426-5500 asia room 1219 chinachem golden plaza 77 mody road tsimshatsui east kowloon hong kong tel: (852) 2721-9778 fax: (852) 2722-1369 japan 9f, tonetsu shinkawa bldg. 1-24-8 shinkawa chuo-ku, tokyo 104-0033 japan tel: (81) 3-3523-3551 fax: (81) 3-3523-7581 memory 2325 orchard parkway san jose, ca 95131, usa tel: 1(408) 441-0311 fax: 1(408) 436-4314 microcontrollers 2325 orchard parkway san jose, ca 95131, usa tel: 1(408) 441-0311 fax: 1(408) 436-4314 la chantrerie bp 70602 44306 nantes cedex 3, france tel: (33) 2-40-18-18-18 fax: (33) 2-40-18-19-60 asic/assp/smart cards zone industrielle 13106 rousset cedex, france tel: (33) 4-42-53-60-00 fax: (33) 4-42-53-60-01 1150 east cheyenne mtn. blvd. colorado springs, co 80906, usa tel: 1(719) 576-3300 fax: 1(719) 540-1759 scottish enterprise technology park maxwell building east kilbride g75 0qr, scotland tel: (44) 1355-803-000 fax: (44) 1355-242-743 rf/automotive theresienstrasse 2 postfach 3535 74025 heilbronn, germany tel: (49) 71-31-67-0 fax: (49) 71-31-67-2340 1150 east cheyenne mtn. blvd. colorado springs, co 80906, usa tel: 1(719) 576-3300 fax: 1(719) 540-1759 biometrics/imagin g/hi-rel mpu/ high speed converters/rf datacom avenue de rochepleine bp 123 38521 saint-egreve cedex, france tel: (33) 4-76-58-30-00 fax: (33) 4-76-58-34-80 literature requests www.atmel.com/literature for more information, please contact: hotline-bdc@gfo.atmel.com 2101d?bdc?06/04 ? atmel corporation 2004 . all rights reserved. atmel ? and combinations thereof, are the registered trade- marks of atmel corporation or its subsidiaries. other terms and product names may be the trademarks of oth- ers.

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